Adaptive matching network

ABSTRACT

A system that incorporates teachings of the present disclosure can include, for example, an apparatus having a matching network adapted to reduce a magnitude of a signal reflection at a port of the matching network. The matching network can have one or more controllable variable reactive elements. A controller can be adapted to determine reflection coefficient information from incident and reflected waves sampled at the port of the matching network, and follow at least one cycle of a coarse tune process for generating one or more control signals to tune one or more reactances of the one or more controllable variable reactive elements. Additional embodiments are disclosed.

PRIOR APPLICATION

This application is a continuation of U.S. patent application Ser. No.13/889,806, filed May 8, 2013, which is a continuation of U.S. patentapplication Ser. No. 12/718,615, filed Mar. 5, 2010 (now U.S. Pat. No.8,463,218), which is a continuation of U.S. patent application Ser. No.11/653,639 filed Jan. 16, 2007 (now U.S. Pat. No. 7,711,337), whichclaims the benefit of Provisional Patent Application Ser. No.60/758,862, filed Jan. 14, 2006 entitled “TECHNIQUES FOR ADAPTIVEIMPEDANCE MATCHING”, by McKinzie et al. All sections of theaforementioned applications are incorporated herein by reference intheir entirety.

BACKGROUND

Mobile communications have become common place throughout society. Notonly is voice communications prevalent, but also the need for mobiledata communications such as email, and Internet browsing has increased.The efficiency of RF systems in mobile communications, such as antennaefficiency of a mobile device as it undergoes changes in itsenvironment, can affect among other things the quality of communicationsexperienced by mobile subscribers. Efficient RF power transfer, or goodimpedance matching, can affect the performance of RF subsystems, such asat the input port of an antenna, between filters, at the output stagesof power amplifiers, and even inter-stage matching between amplifierstages.

Impedance matching networks can be used by various devices and systems,for example, a transmitter, a receiver, a transceiver, a wirelesscommunication station, or a wireless communication device. Examples ofRF systems that can utilize impedance matching networks include withoutlimitation, a wireless Access Point (AP), a modem, a wireless modem, acomputer (e.g., desktop computer, a mobile computer, a laptop computer,a notebook computer, a tablet computer, a server computer, a handheldcomputer, a handheld device, a Personal Digital Assistant (PDA) device,a network, a wireless network, a Local Area Network (LAN), a WirelessLAN (WLAN), a Metropolitan Area Network (MAN), a Wireless MAN (WMAN), aWide Area Network (WAN), a Wireless WAN (WWAN), devices and/or networksoperating in accordance with existing IEEE 802.11x, 802.16x standardsand/or future versions and/or derivatives and/or Long Term Evolution(LTE) of the above standards, a Personal Area Network (PAN), a WirelessPAN (WPAN), units and/or devices which are part of the above WLAN and/orPAN and/or WPAN networks, one way and/or two-way radio communicationsystems, cellular radio-telephone communication systems, a cellulartelephone, a wireless telephone, a Personal Communication Systems (PCS)device, a PDA device which incorporates a wireless communication device,a Multiple Input Multiple Output (MIMO) transceiver or device, a SingleInput Multiple Output (SIMO) transceiver or device, a Multiple InputSingle Output (MISO) transceiver or device, a Multi Receiver Chain (MRC)transceiver or device, a transceiver or device having “smart antenna”technology or multiple antenna technology, or the like.

The above RF systems can utilize any number of RF signaling techniquessuch as, for example, Frequency-Division Multiplexing (FDM), OrthogonalFDM (OFDM), Time-Division Multiplexing (TDM), Time-Division MultipleAccess (TDMA), Extended TDMA (E-TDMA), General Packet Radio Service(GPRS), Extended GPRS, Code-Division Multiple Access (CDMA), WidebandCDMA (WCDMA), CDMA 2000, Multi-Carrier Modulation (MDM), DiscreteMulti-Tone (DMT), Bluetooth®, ZigBee™, or the like.

The above RF systems can utilize impedance matching networks whose loadimpedance can vary with time, temperature, power levels, componentvalues, and many other communication parameters.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 depicts an illustrative embodiment of simultaneous measurementsof magnitude and phase for both forward and reflected waves;

FIG. 2 depicts an illustrative embodiment of a Smith Chart illustratingcourse to fine tuning;

FIG. 3 depicts an illustrative embodiment of simultaneous measurement ofmagnitude and phase for both forward and reflected waves;

FIG. 4 depicts an illustrative embodiment of sequential measurement ofmagnitude for both forward and reflected waves;

FIG. 5 depicts an illustrative embodiment of direct measurement of aratio of the first two nodal voltages;

FIG. 6 depicts an illustrative embodiment of direct measurement of threenodal voltages;

FIG. 7 depicts an illustrative embodiment of an apparatus that can use amethod to compute a terminating impedance of a cascade of 2-portdevices; and

FIG. 8 depicts an illustrative embodiment of a plot of impedance circleson an impedance plane.

DETAILED DESCRIPTION

In the following detailed description, numerous specific details are setforth in order to provide an understanding of the present disclosure.However, it will be understood by those skilled in the art that thepresent disclosure can be practiced without these specific details. Inother instances, well-known methods, procedures, components and circuitshave not been described in detail so as not to obscure the presentdisclosure.

Some portions of the detailed description that follows are presented interms of algorithms and symbolic representations of operations on databits or binary digital signals within a computer memory. Thesealgorithmic descriptions and representations can be the techniques usedby those skilled in the data processing arts to convey the substance oftheir work to others skilled in the art.

An algorithm is here, and generally, considered to be a self-consistentsequence of acts or operations leading to a desired result. Theseinclude physical manipulations of physical quantities. Usually, thoughnot necessarily, these quantities take the form of electrical ormagnetic signals capable of being stored, transferred, combined,compared, and otherwise manipulated. It has proven convenient at times,principally for reasons of common usage, to refer to these signals asbits, values, elements, symbols, characters, terms, numbers or the like.It should be understood, however, that all of these and similar termsare to be associated with the appropriate physical quantities and aremerely convenient labels applied to these quantities.

Unless specifically stated otherwise, as apparent from the followingdiscussions, it is appreciated that throughout the specificationdiscussions utilizing terms such as “processing,” “computing,”“calculating,” “determining,” or the like, refer to the action and/orprocesses of a computer or computing system, or similar electroniccomputing device, that manipulate and/or transform data represented asphysical, such as electronic, quantities within the computing system'sregisters and/or memories into other data similarly represented asphysical quantities within the computing system's memories, registers orother such information storage, transmission or display devices.

Embodiments of the present disclosure can include apparatuses forperforming the operations herein. An apparatus can be speciallyconstructed for the desired purposes, or it can comprise a generalpurpose computing device selectively activated or reconfigured by aprogram stored in the device. Such a program can be stored on a storagemedium, such as, but not limited to, any type of disk including floppydisks, optical disks, compact disc read only memories (CD-ROMs),magnetic-optical disks, read-only memories (ROMs), random accessmemories (RAMs), electrically programmable read-only memories (EPROMs),electrically erasable and programmable read only memories (EEPROMs),magnetic or optical cards, or any other type of media suitable forstoring electronic instructions, and capable of being coupled to asystem bus for a computing device (e.g., non-volatile programmableread-writeable memories such as Flash memories).

The processes and displays presented herein are not inherently relatedto any particular computing device or other apparatus. Various generalpurpose systems can be used with programs in accordance with theteachings herein, or it can prove convenient to construct a morespecialized apparatus to perform the desired method. The desiredstructure for a variety of these systems will appear from thedescription below. In addition, embodiments of the present disclosureare not described with reference to any particular programming language.It will be appreciated that a variety of programming languages can beused to implement the teachings of the present disclosure as describedherein. In addition, it should be understood that operations,capabilities, and features described herein can be implemented with anycombination of hardware (discrete or integrated circuits) and software.

Use of the terms “coupled” and “connected”, along with theirderivatives, can be used. It should be understood that these terms arenot intended as synonyms for each other. Rather, in particularembodiments, “connected” can be used to indicate that two or moreelements are in direct physical or electrical contact with each other.“Coupled” can be used to indicate that two or more elements are ineither direct or indirect (with other intervening elements between them)physical or electrical contact with each other, and/or that the two ormore elements co-operate or interact with each other (e.g., as in acause an effect relationship).

One embodiment of the present disclosure can entail an apparatus havinga matching network adapted to reduce a magnitude of a signal reflectionat a port of the matching network. The matching network can have one ormore controllable variable reactive elements. A controller can beadapted to determine reflection coefficient information from incidentand reflected waves sampled at the port of the matching network, andfollow at least one cycle of a coarse tune process for generating one ormore control signals to tune one or more reactances of the one or morecontrollable variable reactive elements.

One embodiment of the present disclosure can entail a computer-readablestorage medium having computer instructions to determine an inputreflection coefficient from incident and reflected waves detected at aport of a matching network, and coarse tune one or more controllablevariable reactive elements of the matching network with one or morecontrol signals determined according to the input reflectioncoefficient.

One embodiment of the present disclosure can entail a method thatinvolves determining a reflection coefficient from a signal sampled at aport, determining at least one control signal from the reflectioncoefficient, and tuning a matching network with one or more controlsignals, wherein the matching network comprises one or more controllablevariable reactive elements each with an independent control voltage.

Embodiments of the present disclosure can provide several feedbackcontrol system concepts for potential use in an adaptive impedancematching module (AIMM). These concepts can vary in RF system complexity,and hence cost. In an embodiment of the present disclosure, a basictechnical objective can be to minimize the magnitude of the inputreflection coefficient seen at an RF_(in) port under the boundarycondition of a variable load impedance ZL.

Looking at FIG. 1, generally as 100, is a first embodiment usingsimultaneous measurement of magnitude and phase for both forward andreflected waves using identical backward-wave couplers A115 and B 110which sample incident and reflected waves respectively at the input sideof a tuner 120. Coupled and attenuated RF signals can be fed into asingle integrated circuit (IC) which can contain dual channel logamplifiers 127 and 129 followed by gain and phase detectors (such asbuilt into the AD8302 as shown as 125). The dual outputs of the AD8302125 can generate a first voltage, V_(MAG) 135 which is proportional tothe ratio in dB of the input powers (forward and reversed), and a secondvoltage, VPHS 140, which is proportional to the phase difference betweenthe two input signals. These two voltages can be digitally sampled in aclosed loop control system.

The reference plane 145 for the measurement can be approximated asmidway between the two directional couplers 110 and 115, which should belocated as close together as possible. The finite directivity D of thecouplers 110 and 115 sets the minimum detectable reflection coefficient.The two RF paths between the couplers 110 and 115 and the AD8302 125should be as well matched as possible since any differences createmeasurement errors. Also, the frequency response of the couplers 110 and115 should be as close as possible or the differences can be compensatedin software.

The phase detector inside the AD8302 125 can uniquely distinguishinstantaneous phase over a range of only 180°. Thus, the phase can beidentified to within a plus or minus sign. So either Γ or its complexconjugate is known. The tuning algorithm will have to account for thisdegree of uncertainty.

In an embodiment of the present disclosure, a microcontroller or DSPchip 105 can sample the complex reflection coefficient information fromADC1 150 and ADC2 155. Since the reflection coefficient phase angle isknown, a look-up table can be used to immediately perform a coarse tunefunction that feeds approximate bias voltages to the three DACs 160, 165and 170 that in turn control high voltage buffers driving the PTCs 175,180, 185. PTCs are a type of variable reactance network denoted asParascan™ Tunable Capacitors, and they implement a variable capacitorfunction. If the magnitude of the reflection coefficient is not below adesired level, then fine tuning can be accomplished using small anditerative adjustments in bias voltage. Fine tuning can be necessary tocompensate for variations in manufacturing tolerances of tuner componentvalues, or to compensate for temperature variations of the PTCs underhigh power.

In an exemplary embodiment, three PTCs with independent control voltagesare used in the tuner 120. However, it is understood that in general,any finite number of variable reactance networks with independent biasvoltages or bias currents could be included. Also, the exemplaryembodiments herein, a ladder network with series inductors and shuntcaps is described. However, other tuner circuit topologies can also beused, and are thus intended to be within the scope of the presentdisclosure.

As an example to help understand the tuning process, consider the SmithChart shown in FIG. 2 at 200. Assume the initial input reflectioncoefficient at a desired frequency is shown at 215 in this example.Coarse tuning moves the reflection coefficient Γ from point [1] 215 topoint [2] 205 where the magnitude is now |Γ₂|. Application of a finetuning algorithm moves the reflection coefficient from point [2] 205 topoint [3] 210 where the magnitude is |Γ₃|. Repeated application of thefine tuning algorithm decreases Ill further until a specified toleranceis achieved.

The fine tuning algorithm can be a scalar multi-variable minimizationalgorithm where the independent variables are the set of tuning voltagesand the scalar cost function can be the magnitude of the reflectioncoefficient in dB. Many choices exist for this minimization algorithmincluding, but not limited to:

1. Downhill simplex method in multidimensions (section 10.4 of NumericalRecipes);2. Conjugate gradient method in multidimensions (section 10.6 ofNumerical Recipes);3. Quasi-Newton method (section 10.7 of Numerical Recipes).

A digital processor can drive digital-to-analog converters (DACs) whoseoutput voltage is scaled with high voltage buffers to yield PTC biasvoltages of zero to about 30 volts. A charge pump 190 can be used tomultiply a typically available supply voltage of 3.3 volts to more than30 volts to power the voltage buffers, although the present disclosureis not limited in this respect.

The charge pump 335 can be generalized to include any DC-to-DC convertercapable of converting the available supply voltage to a desired higheror lower voltage, and this desired voltage can be positive or negativepolarity, or dual positive and negative polarity. Furthermore, the 30volt maximum PTC voltage used in the above example can be higher orlower depending on the design of the variable capacitors.

The voltage buffers in FIGS. 1 and 3 located between the DACs and PTCscan be replaced with transconductance amplifiers if the PTCs arereplaced with variable reactance networks requiring a bias currentrather than a bias voltage.

Depending on the processor implementation, the ADCs 150 and 155 and DACs160, 165 and 170 can be integrated into the processor IC 105. The meritsof this first embodiment of the present disclosure include that thedigital control system can react very quickly to changes in loadimpedance since coarse tuning can be achieved with only one RFmeasurement. This is possible since both magnitude and phase of thereflection coefficient are simultaneously available.

A second embodiment of the present disclosure is illustrated in FIG. 3at 300 and provides the simultaneous measurement of magnitude for bothforward and reflected waves. In an embodiment of the present disclosure,a single backward-wave coupler 310 can sample incident and reflectedpower at the input side of the tuner 315. Coupled and attenuated RFsignals 305 can be fed into a detector, such as a MAX2016 DualLogarithmic Detector 317. The video output voltages (in dB) can besubtracted internally to create a difference signal at the output OUTD325 which is proportional to the return loss in dB. Measured return lossis given by the simple formula

${{RL}({dB})} = \frac{\left( {V_{OUTD} - V_{CENTER}} \right)}{Slope}$

where V_(CENTER) is the output voltage under the condition of equalvoltages at each input channel. The Slope is about 25 mV/dB. This returnloss can then be digitally sampled in a closed loop control system. Aswith the previous embodiment, the finite directivity D of the couplersets the minimum detectable reflection coefficient.

A microcontroller or DSP chip 320 samples the return loss informationusing ADC1 330. Since the reflection coefficient phase angle is unknown,an iterative tuning algorithm can be required to minimize return loss.The tuning algorithm can be a scalar multi-variable minimization routinewhere the independent variables are the set of tuning voltages and thescalar cost function is the magnitude of the reflection coefficient indB. Many choices exist for this minimization algorithm including:

1. Downhill simplex method in multidimensions (section 10.4 of NumericalRecipes)2. Conjugate gradient method in multidimensions (section 10.6 ofNumerical Recipes)3. Quasi-Newton method (section 10.7 of Numerical Recipes)

The digital processor drives digital-to-analog converters (DACs) 335,340 and 345 whose output voltage is scaled with high voltage buffers toyield PTC bias voltages of zero to about 30 volts. A charge pump 350 canbe used to multiply a typically available supply voltage of 3.3 volts tomore than 30 volts to power the voltage buffers.

Depending on the processor implementation, the ADC 330 and DACs 335, 340and 345 can be integrated into the processor IC 317. The merit of thissecond embodiment is that return loss can be immediately measured in onedigital sample.

Turning now to FIG. 4, is a third embodiment of the present disclosureand provides sequential measurement of magnitude for both forward andreflected waves. In this third embodiment of the present disclosure, aclosed loop control system is built around a low cost MAX4003 logamplifier 425, although the present disclosure is not limited to anyspecific amplifier. A single backward-wave coupler 410 samples incidentand reflected power at the input side of the tuner 415. The incident andreflected power levels are switched at SW₁ 430 such that they can bemeasured in sequence, as controlled by the processor. The MAX4003 426output voltage, which is proportional to coupled power in dB, can bedigitized and the return loss can then be calculated by the processorusing sequential measurements. As with previous embodiments, the finitedirectivity D of the coupler sets the minimum detectable return loss.

The MAX4003 425 log amp was selected because it has a shutdown modewhere it draws only 13 uA of current. Furthermore, when powered, itconsumes only 6 mA from a 3.0 volt supply (18 mW). Again, the presentdisclosure is not limited to using any particular log amp.

Since the microcontroller or DSP chip 420 computes only the return loss(no phase information is available), then an iterative tuning algorithmis required to minimize return loss. The tuning algorithm is a scalarmulti-variable minimization routine where the independent variables arethe set of tuning voltages and the scalar cost function is the magnitudeof the reflection coefficient in dB. Many choices exist for thisminimization algorithm including:

1. Downhill simplex method in multidimensions;2. Conjugate gradient method in multidimensions; and3. Quasi-Newton method.

As with the previous embodiments, the digital processor drivesdigital-to-analog converters (DACs) 435, 440 and 445 whose outputvoltage is scaled with high voltage buffers to yield PTC bias voltagesof zero to about 30 volts. Depending on the processor implementation,the ADC 450 and DACs 435, 440 and 445 can be integrated into theprocessor IC.

The merits of the present third embodiment include, but are not limitedto:

-   -   A relatively low cost log amp is employed. At the time of the        present disclosure, the MAX4003 sells for ˜$1.09 in qty of 100.    -   The MAX4003 log amp consumes only 18 mW of power during normal        operation at 3.0 volts.    -   The log amp can be powered down when power measurements are not        required.

Turning now to FIG. 5, is a fourth embodiment of the present disclosureand provides direct measurement of the ratio of the first two nodalvoltages. This embodiment is designed to offer an “indirect” measurementof input impedance or input reflection coefficient for the tuner 510. Incontrast, a direct measurement would involve directional couplers as inprevious embodiments. By eliminating the directional couplers one savesBill of Material (BOM) cost and board real estate and eliminates abandwidth restriction caused by miniature narrowband couplers.

The input impedance sensing circuit consists of two additional knownreactive components on the input side of the tuner, namely Y_(m1) 535and Z_(m2)=1/Y_(m2) 540. RF voltages V1 and V2 are measured using highimpedance (relative to Zo=50 W) resistive voltage dividers. The inputimpedance can be expressed as

$\begin{matrix}{Z_{in} = \frac{V_{1}}{I_{in}}} \\{= \frac{V_{1}}{{V_{1}Y_{m\; 1}} + \left( \frac{V_{1} - V_{2}}{Z_{m\; 2}} \right)}} \\{= \frac{1}{Y_{m\; 1} + \left( \frac{1 - \frac{V_{2}}{V_{1}}}{Z_{m\; 2}} \right)}} \\{= {\frac{1}{Y_{in}}.}}\end{matrix}$

Since the input reflection coefficient Γ can be expressed in terms ofinput admittance, then

$\begin{matrix}{\Gamma = \frac{Y_{o} - Y_{in}}{Y_{o} + Y_{in}}} \\{= \frac{Y_{o} - Y_{m\; 1} - \left( \frac{1 - \frac{V_{2}}{V_{1}}}{Z_{m\; 2}} \right)}{Y_{o} + Y_{m\; 1} + \left( \frac{1 - \frac{V_{2}}{V_{1}}}{Z_{m\; 2}} \right)}} \\{= {\frac{Y_{o} - Y_{m\; 1} - {Y_{m\; 2}\left( {1 - \frac{V_{2}}{V_{1}}} \right)}}{Y_{o} + Y_{m\; 1} + {Y_{m\; 2}\left( {1 - \frac{V_{2}}{V_{1}}} \right)}}.}}\end{matrix}$

Hence the complex value of Γ is known with one digital sample of thecomplex ratio of nodal voltages. It should be noted that componentsY_(m1) 535 and Z_(m2) 540 are not restricted, but they must be known.Their values are chosen by the system designer, and Y_(m1) 535 can beset to zero (omitted) if desired. Only a series component is requiredfor this approach to work. The accuracy of the indirectly measured Γ isdefined largely by the component tolerances of Y_(m1) 535 and Z_(m2)540.

One could design the tuner 510 such that Y_(m1) 535 is the first shuntvoltage tunable capacitor (PTC) and Z_(m2) 540 is the first seriesinductor, or a short series transmission line. However, this wouldrequire that the PTC capacitance be known very accurately for all biasvoltages and temperatures. While it is conceivable to obtain suchdetailed information, it cannot be practical in high volume productiondepending on the tolerance required

A microcontroller or DSP chip 530 samples the complex node voltage ratiofrom ADC1 545 and ADC2 550 and calculates the complex input reflectioncoefficient directly from the equation above. A look-up table can beused to immediately perform a coarse tune function that feedsapproximate bias voltages to the three DACs 555, 560 and 565 that inturn control high voltage buffers for the PTCs 515, 520, 525. If themagnitude of the reflection coefficient is not below a desired level,then fine tuning can be accomplished using small and iterativeadjustments in bias voltage. Fine tuning can be necessary to compensatefor variations in manufacturing tolerances of tuner component values, orto compensate for temperature variations of the PTCs under high power.

The fine tuning algorithm can be a scalar multi-variable minimizationalgorithm where the independent variables are the set of tuning voltagesand the scalar cost function can be the magnitude of the reflectioncoefficient in dB. Many choices exist for this minimization algorithmincluding

1. Downhill simplex method in multidimensions;2. Conjugate gradient method in multidimensions; and3. Quasi-Newton method.

The digital processor 530 drives digital-to-analog converters (DACs)555, 560 and 565 whose output voltage is scaled with high voltagebuffers to yield PTC bias voltages of zero to about 30 volts. A chargepump 570 can be used to multiply a typically available supply voltage of3.3 volts to more than 30 volts to power the voltage buffers. Dependingon the processor implementation, the ADCs 545 and 550 and DACs 555, 560and 565 can be integrated into the processor IC.

The merits of the present embodiment shown in FIG. 5 include:

-   -   Board real estate can be reduced significantly because        directional couplers are not needed, and the resistive dividers        occupy a very small footprint.    -   The cost of directional couplers is eliminated.    -   The bandwidth of the reflection coefficient sensing circuit is        significantly increased relative to using miniature ceramic        hybrid couplers.    -   The digital control system can react very quickly to changes in        load impedance since course tuning can be achieved with only one        RF measurement. This is possible since both magnitude and phase        of the first two nodal voltages are simultaneously available.

Turning now to FIG. 6, is a fifth embodiment of the present disclosureand provides direct measurement of the ratio of the first two nodalvoltages. In this embodiment is a modification of embodiment 4 wherethree node voltages are measured instead of two, and only theirmagnitudes are measured using a single channel log amp or temperaturecompensated diode detector. Ratios of node voltages are calculated bythe microcontroller/DSP 640. Any ambiguity of V1 and V2 used tocalculate Z1 based on magnitude measurements can be resolved bycalculating Z2 from a measurement of a second pair of voltages, V2 andV3. Then Z2 is mapped into Z1 given the known values of shunt and seriesmeasurement impedances. In this manner, three measurements of nodevoltage magnitude permit a unique determination of the input impedanceZ1 for the tuner.

The first through third embodiments described above can use directionalcouplers to measure forward and reflected power. So for theseembodiments, the minimum dynamic range needed by the detector is themagnitude of the best case return loss that is desired to be resolved,plus the dynamic range of the input RF signal. So if it is desired toresolve a return loss down to −20 dB and operate the AIMM over a 30 dBdynamic range of input powers, then a 50 dB (20 dB+30 dB) log amplifiercan be needed. In contrast, the fourth and fifth embodiments measure thetotal RF voltage at the nodes. These voltages are expected to be fairlysimilar in magnitude, especially for a well matched tuner. So thedetector's required dynamic range is expected to be less for embodiments4 and 5.

Current consumption will also be less for the MAX2205-2208 family ofdetectors relative to a log amp. They typically consume only 3.5 mA orless at 3 volts, and 0.5 uA at shutdown. The ability to create asuccessful AIMM depends on two critical technical achievements. Thefirst requirement is to create a highly-linear, series network of lowloss, tunable capacitors. But the second requirement is for amonolithic, low cost, logarithmic amplifier with a broad dynamic range.Dynamic range is very important for many cell phone applications wheretransmit power control over multiple decades is required, although thepresent disclosure is not limited in this respect.

The advent of a log amp with an integrated phase detector provides adramatic advantage in closed loop settling time compared to conventionallog amps with only envelope detection. The reason for the advantage isthat phase and magnitude information are used together to achieve coarsetuning with only one sample of reflection coefficient or node voltage.The only commercially available log amp with a phase detector is AnalogDevices part number AD8302. However, the cost of the AD8302 is expectedto be an order of magnitude higher than a conventional single channellog amp. One of the major drawbacks of the AD8302 is its relatively highcurrent consumption at 20 mA and a shutdown feature is needed on afuture version of this part. As with FIG. 5, switch SW1 is shown at 645and tuner 610 can include voltage tunable capacitors, such as voltagetunable dielectric varactors, which can be referred to as Parascan®Tunable Capacitors (PTCs). Charge pump 630 can also be included such aswith the charge pump of FIG. 5.

In some embodiments of the present disclosure described above, theimpedances added to the tuner for measurements of Γ in the fourthembodiment can be any reactance. However, an obvious option is to use ashunt capacitor followed by a series inductor. This will preserve theladder circuit topology that was employed in each of the previousembodiments.

Looking now at FIG. 7 is an embodiment of the present disclosure thatillustrates a method to compute the terminating impedance of a cascadeof 2-port devices 700 which are characterized through transmission (orABCD) parameters and to which a signal from a source with a knownimpedance is applied by measuring the magnitude of the voltages at theinput and output of the cascade and between the devices. Depicted inFIG. 7 is:

-   -   source voltage U_(s) 705; reference impedance R, 710; network        elements Z_(m) 725 and 740 and Y_(m) 720 and 735; terminating        impedance Z_(r) 745; input voltage U_(i) 715; voltage U_(c) 730;        and output voltage U_(o) 750,

Indices i := 0 c := 1 o := 2 Source Voltage Us := 2 V ReferenceImpedance Rw := 50 Ω Network Elements Zm := (1 + 4.5j 1 − 8j)^(T) Ym :=(0.2 − 0.4j 0.075 − 0.264j)^(T)|ho Terminating Impedance Zt := 20 − 75jΩ

Given this information we can compute the input voltage U_(i) as

$U_{i}:={{{\frac{1}{1 + {\left( {Y\; m_{0}}\rightarrow\frac{1}{{Z\; m_{0}} + \frac{1}{{Ym}_{1} + \frac{1}{{Zm}_{1} + {Zt}}}} \right) \cdot {Rw}}} \cdot {Us}}{U_{i}}} = 0.069824}$

the voltage Uc as

$U_{c}:={\frac{\frac{1}{\left. {Ym}_{1}\leftarrow\frac{1}{{Zm}_{1} + {Zt}} \right.}}{{Z\; m_{0}} + \frac{1}{{Ym}_{1} + \frac{1}{{Zm}_{1} + {Zt}}}} \cdot U_{i}}$U_(c) = 0.031494

and the output voltage Uo as

$U_{o}:={\frac{Zt}{{Zm}_{1} + {Zt}} \cdot U_{c}}$ U_(o) = 0.028553

Transmission parameters

$\begin{matrix}\begin{matrix}{{A\; 1}:={\begin{pmatrix}\frac{1}{Z\; m_{0}} & 1 \\\frac{{Ym}_{0}}{{Zm}_{0}} & {{Ym}_{0} + \frac{1}{{Zm}_{0}}}\end{pmatrix} \cdot {Zm}_{0}}} & {{A\; 1} = \begin{pmatrix}1 & {1 + {4.5\; j}} \\{0.2 - {0.4\; j}} & {3 + {0.5\; j}}\end{pmatrix}} \\{{A\; 2}:={\begin{pmatrix}\frac{1}{\; {Zm}_{1}} & 1 \\\frac{{Ym}_{1}}{{Zm}_{1}} & {{Ym}_{1} + \frac{1}{{Zm}_{1}}}\end{pmatrix} \cdot {Zm}_{1}}} & {{A\; 2} = \begin{pmatrix}1 & {1 - {8\; j}} \\\begin{matrix}{0.075 -} \\{0.264\; j}\end{matrix} & \begin{matrix}{{- 1.037} -} \\{0.864\; j}\end{matrix}\end{pmatrix}} \\{{Ac}:={A\; {1 \cdot A}\; 2}} & {{Ac} = \begin{pmatrix}\begin{matrix}{2.263 +} \\{0.073\; j}\end{matrix} & \begin{matrix}{3.851 -} \\{13.531\; j}\end{matrix} \\\begin{matrix}{0.557 -} \\{1.155\; j}\end{matrix} & \begin{matrix}{{- 5.679} -} \\{5.111\; j}\end{matrix}\end{pmatrix}}\end{matrix} & \; \\{\mspace{79mu} \begin{matrix}{U_{i} = {\frac{Z_{i}}{{Rw} + Z_{i}} \cdot {Us}}} \\{= \frac{\left( {{{Ac}_{0,0} \cdot {Zt}} + {Ac}_{0,1}} \right) \cdot {Us}}{\begin{matrix}{{\left( {{{Ac}_{1,0} \cdot {Zt}} + {Ac}_{1,1}} \right) \cdot {Rw}} +} \\{{{Ac}_{0,0} \cdot {Zt}} + {Ac}_{0,1}}\end{matrix}}}\end{matrix}} & (1) \\\begin{matrix}{\mspace{79mu} {U_{o} = \frac{{Zt} \cdot U_{i}}{{{Ac}_{0,0} \cdot {Zt}} + {Ac}_{0,1}}}} \\{= \frac{{Zt} \cdot {Us}}{\begin{matrix}{{\left( {{{Ac}_{1,0} \cdot {Zt}} + {Ac}_{1,1}} \right) \cdot {Rw}} +} \\{{{Ac}_{0,0} \cdot {Zt}} + {Ac}_{0,1}}\end{matrix}}}\end{matrix} & (2) \\{\mspace{79mu} {U_{c} = \frac{\left( {{A\; {2_{0,0} \cdot {Zt}}} + {A\; 2_{0,1}}} \right) \cdot {Us}}{\begin{matrix}{{\left( {{{Ac}_{1,0} \cdot {Zt}} + {Ac}_{1,1}} \right) \cdot {Rw}} +} \\{{{Ac}_{0,0} \cdot {Zt}} + {Ac}_{0,1}}\end{matrix}}}} & (3)\end{matrix}$

We divide the transmission parameters and the termination into real andimaginary components

Ar1:=Re(A1) Ai1:=Im(A1)

Ar2:=Re(A2) Ai2:=Im(A2)

Arc:=Re(Ac) Aic:=Im(Ac)

Xt:=Re(Zt) Yt:=Im(Zt)

and express the magnitudes of the measured voltages as

$\begin{matrix}{\left( {U_{i}} \right)^{2} = \frac{\begin{bmatrix}{\left( {{{Arc}_{0,0} \cdot {Xt}} - {{Aic}_{0,0} \cdot {Yt}} + {Arc}_{0,1}} \right)^{2} +} \\\left( {{{Aic}_{0,0} \cdot {Xt}} - {{Arc}_{0,0} \cdot {Yt}} + {Aic}_{0,1}} \right)^{2}\end{bmatrix} \cdot {Us}^{2}}{\begin{matrix}{{\begin{bmatrix}{{\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot {Xt}} -} \\{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot} \\{{Yt} + {{Arc}_{1,1} \cdot {Rw}} + {Arc}_{0,1}}\end{bmatrix}^{2}\ldots} +} \\\left\lceil \begin{matrix}{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot} \\{{Xt} + {\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot}} \\{{Yt} + {{Aic}_{1,1} \cdot {Rw}} + {Aic}_{0,1}}\end{matrix} \right\rceil^{2}\end{matrix}}} & (4) \\{\left( {U_{c}} \right)^{2} = \frac{\begin{bmatrix}{\left( {{{Ar}\; {2_{0,0} \cdot {Xt}}} - {{Ai}\; {2_{0,0} \cdot {Yt}}} + {{Ar}\; 2_{0,1}}} \right)^{2} +} \\\left( {{{Ai}\; {2_{0,0} \cdot {Xt}}} + {{Ar}\; {2_{0,0} \cdot {Yt}}} + {{Ai}\; 2_{0,1}}} \right)^{2}\end{bmatrix} \cdot {Us}^{2}}{\begin{matrix}{{\begin{bmatrix}{{\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot {Xt}} -} \\{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot} \\{{Yt} + {{Arc}_{1,1} \cdot {Rw}} + {Arc}_{0,1}}\end{bmatrix}^{2}\ldots} +} \\\left\lceil \begin{matrix}{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot} \\{{Xt} + {\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot}} \\{{Yt} + {{Aic}_{1,1} \cdot {Rw}} + {Aic}_{0,1}}\end{matrix} \right\rceil^{2}\end{matrix}}} & (5) \\{\left( {U_{o}} \right)^{2} = \frac{\left( {{Xt}^{2} + {Yt}^{2}} \right) \cdot {Us}^{2}}{\begin{matrix}{{\begin{bmatrix}{{\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot {Xt}} -} \\{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot} \\{{Yt} + {{Arc}_{1,1} \cdot {Rw}} + {Arc}_{0,1}}\end{bmatrix}^{2}\ldots} +} \\\begin{bmatrix}{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot} \\{{Xt} + {\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot}} \\{{Yt} + {{Aic}_{1,1} \cdot {Rw}} + {Aic}_{0,1}}\end{bmatrix}^{2}\end{matrix}}} & (6)\end{matrix}$

We solve (6) for Us²

$\begin{matrix}{{Us}^{2} = \frac{\begin{bmatrix}{{\begin{bmatrix}{{\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot {Xt}} -} \\{{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot {Yt}} +} \\{{{Arc}_{1,1} \cdot {Rw}} + {Arc}_{0,1}}\end{bmatrix}^{2}\ldots} +} \\\left\lceil \begin{matrix}{{\left( {{{Aic}_{1,0} \cdot {Rw}} + {Aic}_{0,0}} \right) \cdot {Xt}} +} \\{\left( {{{Arc}_{1,0} \cdot {Rw}} + {Arc}_{0,0}} \right) \cdot} \\{{Yt} + {{Aic}_{1,1} \cdot {Rw}} + {Aic}_{0,1}}\end{matrix} \right\rceil^{2}\end{bmatrix} \cdot \left( {U_{o}} \right)^{2}}{\left( {{Xt}^{2} + {Yt}^{2}} \right)}} & \left( {6a} \right)\end{matrix}$

and substitute (6a) in (4) and (5)

$\begin{matrix}{\left( {U_{i}} \right)^{2} = \frac{\left\lceil \begin{matrix}{\left( {{{Arc}_{0,0} \cdot X_{t}} - {{Aic}_{0,0} \cdot {Yt}} + {Arc}_{0,1}} \right)^{2} +} \\\left( {{{Aic}_{0,0} \cdot {Xt}} + {{Arc}_{0,0} \cdot {Yt}} + {Aic}_{0,1}} \right)^{2}\end{matrix} \right\rceil \cdot \left( {U_{o}} \right)^{2}}{{Xt}^{2} + {Yt}^{2}}} & (7) \\{\left( {U_{c}} \right)^{2} = \frac{\left\lceil \begin{matrix}{\left( {{{Ar}\; {2_{0,0} \cdot X_{t}}} - {{Ai}\; {2_{0,0} \cdot {Yt}}} + {{Ar}\; 2_{0,1}}} \right)^{2} +} \\\left( {{{Ai}\; {2_{0,0} \cdot {Xt}}} + {{Ar}\; {2_{0,0} \cdot {Yt}}} + {{Ai}\; 2_{0,1}}} \right)^{2}\end{matrix} \right\rceil \cdot \left( {U_{o}} \right)^{2}}{{Xt}^{2} + {Yt}^{2}}} & (8)\end{matrix}$

(7) and (8) can be written in the form

$\begin{matrix}{{\left\lbrack {{Xt} - \frac{{Re}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)}{\left( \frac{U_{i}}{U_{o}} \right)^{2} - \left( {{Ac}_{0,0}} \right)^{2}}} \right\rbrack^{2} + \left\lbrack {{Yt} + \frac{{Im}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)}{\left( \frac{U_{i}}{U_{o}} \right)^{2} - \left( {{Ac}_{0,0}} \right)^{2}}} \right\rbrack^{2}} = \left\lbrack \frac{{{Ac}_{0,1}} \cdot \frac{U_{i}}{U_{0}}}{\left( \frac{U_{i}}{U_{o}} \right)^{2} - \left( {{Ac}_{0,0}} \right)^{2}} \right\rbrack^{2}} & (9) \\{{\left\lbrack {{Xt} - \frac{{Re}\left( {{Ac}_{0,0} \cdot \left( \overset{\_}{A\; 2_{0,1}} \right)} \right.}{\left( \frac{U_{c}}{U_{o}} \right)^{2} - \left( {{A\; 2_{0,0}}} \right)^{2}}} \right\rbrack^{2} + \left\lbrack {{Yt} + \frac{{Im}\left( {A\; {2_{0,0} \cdot \overset{\_}{A\; 2_{0,1}}}} \right)}{\left( \frac{U_{c}}{U_{o}} \right)^{2} - \left( {{A\; 2_{0,0}}} \right)^{2}}} \right\rbrack^{2}} = \left\lbrack \frac{{{A\; 2_{0,1}}} \cdot \frac{U_{c}}{U_{o}}}{\left( \frac{U_{c}}{U_{o}} \right)^{2} - \left( {{A\; 2_{0,0}}} \right)^{2}} \right\rbrack^{2}} & (10)\end{matrix}$

We also solve (4) for Us² and substitute in (5), resulting in

$\begin{matrix}{\left( {U_{c}} \right)^{2} = {\frac{\begin{bmatrix}{\left( {{{Ar}\; {2_{0,0} \cdot {Xt}}} - {{Ai}\; {2_{0,0} \cdot {Yt}}} + {{Ar}\; 2_{0,1}}} \right)^{2} +} \\\left( {{{Ai}\; {2_{0,0} \cdot {Xt}}} + {{Ar}\; {2_{0,0} \cdot {Yt}}} + {{Ai}\; 2_{0,1}}} \right)^{2}\end{bmatrix}}{\begin{bmatrix}{\left( {{{Ar}\; {c_{0,0} \cdot {Xt}}} - {{Ai}\; {c_{0,0} \cdot {Yt}}} + {{Ar}\; c_{0,1}}} \right)^{2} +} \\\left( {{{Ai}\; {c_{0,0} \cdot {Xt}}} + {{Ar}\; {c_{0,0} \cdot {Yt}}} + {{Ai}\; c_{0,1}}} \right)^{2}\end{bmatrix}} \cdot \left( {U_{i}} \right)^{2}}} & (11)\end{matrix}$

from which we derive

$\begin{matrix}{{{\left\lbrack {{Xt} + \frac{\begin{matrix}{{{{{Re}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}}\mspace{14mu} \ldots} +} \\{- {{Re}\left( {A\; {2_{0,0} \cdot \overset{\_}{A\; 2_{0,1}}}} \right)}}\end{matrix}}{{\left( {{Ac}_{0,0}} \right)^{2} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - \left( {{A\; 2_{0,0}}} \right)^{2}}} \right\rbrack^{2}\mspace{14mu} \ldots} = {\frac{\begin{bmatrix}\begin{matrix}{{\left( {{{Ac}_{0,0}} \cdot {{A\; 2_{0,1}}}} \right)^{2}\mspace{14mu} \ldots} +} \\{{\left( {{{Ac}_{0,1}} \cdot {{A\; 2_{0,0}}}} \right)^{2}\mspace{14mu} \ldots} +}\end{matrix} \\{{- 2} \cdot {{Re}\left( {{{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}} \cdot \overset{\_}{A\; 2_{0,0}} \cdot A}\; 2_{0,1}} \right)}}\end{bmatrix} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}}{\left\lbrack {{\left( {{Ac}_{0,0}} \right)^{2} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - \left( {{A\; 2_{0,0}}} \right)^{2}} \right\rbrack^{2}} + \left\lbrack {{Yt} - \frac{\begin{matrix}{{{{{Im}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}}\mspace{14mu} \ldots} +} \\{{Im}\left( {A\; {2_{0,0} \cdot \overset{\_}{A\; 2_{0,1}}}} \right)}\end{matrix}}{{\left( {{Ac}_{0,0}} \right)^{2} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - \left( {{A\; 2_{0,0}}} \right)^{2}}} \right\rbrack^{2}}}\mspace{14mu}} & (12)\end{matrix}$

(9), (10) and (12) are in the form

(Xt−X)²+(Yt−Y)² =R ²

and so constitute circles on the impedance plane

$\begin{matrix}{X_{i}:=\frac{{Re}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)}{\left( \frac{U_{i}}{U_{o}} \right)^{2} - \left( {{A\; c_{0,0}}} \right)^{2}}} & {X_{c}:=\frac{{Re}\left( {A\; {2_{0,0} \cdot \overset{\_}{A\; 2_{0,1}}}} \right)}{\left( \frac{U_{c}}{U_{o}} \right)^{2} - \left( {{A\; 2_{0,0}}} \right)^{2}}} \\{Y_{i}:=\frac{{Im}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)}{\left( \frac{U_{i}}{U_{o}} \right)^{2} - \left( {{A\; c_{0,0}}} \right)^{2}}} & {Y_{c}:=\frac{{Im}\left( {A\; {2_{0,0} \cdot \overset{\_}{A\; 2_{0,1}}}} \right)}{\left( \frac{U_{c}}{U_{o}} \right)^{2} - \left( {{A\; 2_{0,0}}} \right)^{2}}} \\{R_{i}:={\frac{{{Ac}_{0,1}} \cdot \frac{U_{i}}{U_{o}}}{\left( \frac{U_{i}}{U_{o}} \right)^{2} - \left( {{Ac}_{0,0}} \right)^{2}}}} & {R_{c}:={\frac{{{A\; 2_{0,1}}} \cdot \frac{U_{c}}{U_{o}}}{\left( \frac{U_{c}}{U_{o}} \right)^{2} - \left( {{A\; 2_{0,0}}} \right)^{2}}}}\end{matrix}$$X_{o}:=\frac{{{{Re}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - {{Re}\left( {A\; {2_{0,0} \cdot \overset{\_}{A\; 2_{0,1}}}} \right)}}{{\left( {{A\; c_{0,0}}} \right)^{2} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - \left( {{A\; 2_{0,0}}} \right)^{2}}$$Y_{o}:=\frac{{{{Im}\left( {{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}}} \right)} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - {{Im}\left( {A\; {2_{0,0} \cdot \overset{\_}{A\; 2_{0,1}}}} \right)}}{{\left( {{A\; c_{0,0}}} \right)^{2} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - \left( {{A\; 2_{0,0}}} \right)^{2}}$$R_{o}:={\frac{\sqrt{\begin{matrix}{\left( {{{Ac}_{0,0}} \cdot {{A\; 2_{0,1}}}} \right)^{2} + {\left( {{{Ac}_{0,1}} \cdot {{A\; 2_{0,0}}}} \right)^{2}\mspace{14mu} \ldots} +} \\{{- 2} \cdot {{Re}\left( {{{Ac}_{0,0} \cdot \overset{\_}{{Ac}_{0,1}} \cdot \overset{\_}{A\; 2_{0,0}} \cdot A}\; 2_{0,1}} \right)}}\end{matrix}} \cdot \frac{U_{c}}{U_{i}}}{{\left( {{Ac}_{0,0}} \right)^{2} \cdot \left( \frac{U_{c}}{U_{i}} \right)^{2}} - \left( {{A\; 2_{0,0}}} \right)^{2}}}$

The 2 circles intersect in 2 points, one of which represents theterminating impedance. The functions functions are useful to plot theimpedance plane circles and to find the intersections of 2 circles.

Functions to plot circles

${{yc}\left( {x,{xo},{yo},r} \right)}:=\begin{bmatrix}{{yo} + \sqrt{r^{2} - \left( {x - {xo}} \right)^{2}}} \\{{yo} - \sqrt{r^{2} - \left( {x - {xo}} \right)^{2}}}\end{bmatrix}$

Find real components of intersections

${{xcint}\left( {x,y,r} \right)}:={\frac{\begin{bmatrix}{{{\left\lbrack {\left( {x_{1} - x_{0}} \right)^{2} + \left( {y_{1} - y_{0}} \right)^{2} + \left( r_{0} \right)^{2} - \left( r_{1} \right)^{2}} \right\rbrack \cdot \left( {x_{1} - x_{0}} \right)}\mspace{14mu} \ldots} +} \\{\left( {y_{1} - y_{0}} \right) \cdot \sqrt{\begin{matrix}{\left( {2 \cdot r_{0} \cdot r_{1}} \right)^{2} -} \\\begin{bmatrix}{\left( {x_{1} - x_{0}} \right)^{2} +} \\{\left( {y_{1} - y_{0}} \right)^{2} - \left( r_{1} \right)^{2} - \left( r_{0} \right)^{2}}\end{bmatrix}^{2}\end{matrix}}} \\{{{\left\lceil {\left( {x_{1} - x_{0}} \right)^{2} + \left( {y_{1} - y_{0}} \right)^{2} + \left( r_{0} \right)^{2} - \left( r_{1} \right)^{2}} \right\rceil \cdot \left( {x_{1} - x_{0}} \right)}\mspace{14mu} \ldots} +} \\{{- \left( {y_{1} - y_{0}} \right)} \cdot \sqrt{\begin{matrix}{\left( {2 \cdot r_{0} \cdot r_{1}} \right)^{2} -} \\\begin{bmatrix}{\left( {x_{1} - x_{0}} \right)^{2} + \left( {y_{1} - y_{0}} \right)^{2} -} \\{\left( r_{1} \right)^{2} - \left( r_{0} \right)^{2}}\end{bmatrix}^{2}\end{matrix}}}\end{bmatrix}}{2\left\lbrack {\left( {x_{1} - x_{0}} \right)^{2} + \left( {y_{1} - y_{0}} \right)^{2}} \right\rbrack} + x_{0}}$

Find both intersections on Z-plane representing 2 possible solutions

${{CircInt}\left( {x,y,r} \right)}:={\begin{matrix}\left. ɛ\leftarrow{\sqrt{r_{0} \cdot r_{1}} \cdot 10^{- 10}} \right. \\\left. C\leftarrow{{xcint}\left( {x,y,r} \right)} \right. \\{{{for}\mspace{14mu} i} \in {0\mspace{14mu} \ldots \mspace{14mu} 1}} \\{\begin{matrix}\begin{matrix}{{{for}\mspace{14mu} k} \in {0\mspace{14mu} \ldots \mspace{14mu} 1}} \\\left. Y^{(k)}\leftarrow{{yc}\left( {C_{i},x_{k},y_{k},r_{k}} \right)} \right. \\{\left. m\leftarrow \right.\begin{matrix}0 & {{{if}\; \left( {{{Y_{0,0} - Y_{0,1}}} < ɛ} \right)} + \left( {{{Y_{0,0} - Y_{1,1}}} < ɛ} \right)} \\1 & {otherwise}\end{matrix}}\end{matrix} \\\left. {CI}_{i}\leftarrow{C_{i} + {j \cdot Y_{m,0}}} \right.\end{matrix}} \\{CI}\end{matrix}}$

Looking now at FIG. 8 at 800:

We plot the circles on the impedance plane

Plot formatting

Number of samples Ns := 2001 U_(i) circle Xmin_(i) := X_(i) − R_(i)Xmax_(i) := X_(i) + R_(i)${\Delta \; x_{i}}:=\frac{{Xmax}_{i} - {Xmin}_{i}}{1.01 - \left( {{Ns} - 1} \right)}$xi := Xmin_(i), Xmin_(i) + Δx_(i)..Xmax_(i) U_(c) cicle Xmin_(c) :=X_(c) − R_(c) Xmax_(c) := X_(c) + R_(c)${\Delta \; x_{c}}:=\frac{{Xmax}_{c} - {Xmin}_{c}}{{Ns} - 1}$ xc :=Xmin_(c), Xmin_(c) + Δx_(c)..Xmax_(c) U_(o) circle Xmin_(o) := X_(o) −R_(o) Xmax_(o) := X_(o) + R_(o)${\Delta x}_{o}:=\frac{{Xmax}_{o} - {Xmin}_{o}}{{Ns} - 1}$ xo :=Xmin_(o), Xmin_(o) + Δx_(o)..Xmax_(o) Intersections${{CircInt}\left( {X,Y,R} \right)} = \begin{pmatrix}{7.036 + {4.05j}} \\{20 - {75j}}\end{pmatrix}$ Actual impedance Zt = 20 − 75j

Given Zt, the input impedance, or the load seen by source

${Zi} = \frac{U_{i}}{I_{i}}$

can be computed as

${Zi}:=\frac{{{Ac}_{0,0} \cdot {Zt}} + {Ac}_{0,1}}{{{Ac}_{1,0} \cdot {Zt}} + {Ac}_{1,1}}$Zi = 0.722 + 1.618 j

This car be verified by direct computation from the network elements as

$\frac{1}{{Ym}_{0} + \frac{1}{{Zm}_{0} + \frac{1}{{Ym}_{1} + \frac{1}{{Zm}_{1}} + {Zt}}}} = {0.722 + {1.618\; j}}$

The variable reactive elements referred to above can be variablecapacitances, variable inductances, or both. The variable capacitors canbe semiconductor varactors, microelectromechanical system (MEMS)varactors, MEMS switched capacitors, and/or voltage tunable dielectriccapacitors—although the present invention is not limited in thisrespect.

Some embodiments of the present disclosure can be implemented, forexample, using a machine-readable medium or article which can store aninstruction or a set of instructions that, if executed by a machine, forexample, by a system of the present disclosure which includes abovereferenced controllers and DSPs, or by other suitable machines, causethe machine to perform a method and/or operations in accordance withembodiments of the present disclosure. Such machine can include, forexample, any suitable processing platform, computing platform, computingdevice, processing device, computing system, processing system,computer, processor, or the like, and can be implemented using anysuitable combination of hardware and/or software.

The machine-readable medium or article can include, for example, anysuitable type of memory unit, memory device, memory article, memorymedium, storage device, storage article, storage medium and/or storageunit, for example, memory, removable or non-removable media, erasable ornon-erasable media, writeable or re-writeable media, digital or analogmedia, hard disk, floppy disk, Compact Disk Read Only Memory (CD-ROM),Compact Disk Recordable (CD-R), Compact Disk Re-Writeable (CD-RW),optical disk, magnetic media, various types of Digital Versatile Disks(DVDs), a tape, a cassette, or the like. The instructions can includeany suitable type of code, for example, source code, compiled code,interpreted code, executable code, static code, dynamic code, or thelike, and can be implemented using any suitable high-level, low-level,object-oriented, visual, compiled and/or interpreted programminglanguage, e.g., C, C++, Java, BASIC, Pascal, Fortran, Cobol, assemblylanguage, machine code, or the like.

An embodiment of the present disclosure provides a machine-accessiblemedium that provides instructions, which when accessed, cause a machineto perform operations comprising minimizing the magnitude of an inputreflection coefficient seen at an RFin port under boundary conditions ofa variable load impedance ZL by an adaptive antenna impedance matchingmodule (AIMM) by using a tuner connected to said AIMM and including aplurality of voltage tunable capacitors with independent controlvoltages within said tuner, wherein backward-wave couplers sampleincident and reflected waves respectively at the input side of saidtuner; and using a microcontroller or digital signal process (DSP) chipto sample complex reflection coefficient information from said incidentand reflected waves and providing by said microcontroller or DSP acoarse tune function that feeds approximate bias voltages to controlsaid voltage tunable capacitors. The machine-accessible medium canfurther comprise the instructions causing the machine to performoperations further comprising sampling the complex reflectioncoefficient information from at least one analog to digital converter(ADC) by said microcontroller or DSP chip.

Some embodiments of the present disclosure can be implemented bysoftware, by hardware, or by any combination of software and/or hardwareas can be suitable for specific applications or in accordance withspecific design requirements. Embodiments of the present disclosure caninclude units and/or sub-units, which can be separate of each other orcombined together, in whole or in part, and can be implemented usingspecific, multi-purpose or general processors or controllers, or devicesas are known in the art. Some embodiments of the present disclosure caninclude buffers, registers, stacks, storage units and/or memory units,for temporary or long-term storage of data or in order to facilitate theoperation of a specific embodiment.

While the present disclosure has been described in terms of what are atpresent believed to be its preferred embodiments, those skilled in theart will recognize that various modifications to the discloseembodiments can be made without departing from the scope of the presentdisclosure as defined by the following claims.

What is claimed is:
 1. A communication device, comprising: an antenna; atransceiver coupled with the antenna; an RF coupler; a matching networkcoupled with the antenna, the transceiver and the RF coupler, thematching network comprising a variable reactance element; and a controlcircuit coupled with the matching network and the RF coupler, whereinthe control circuit obtains measurements via the RF coupler for forwardand reverse waves, wherein the control circuit performs an impedancetuning process by adjusting a reactance of the variable reactanceelement according to the measurements to reduce a return loss.
 2. Thecommunication device of claim 1, wherein the RF coupler comprises adirectional coupler connected between the matching network and thetransceiver that samples the forward and reverse waves at a port of thematching network.
 3. The communication device of claim 1, wherein thecontrol circuit determines reflection coefficient information fromsamples provided by an analog to digital converter based on themeasurements.
 4. The communication device of claim 1, wherein a look-uptable is used to perform the impedance tuning process.
 5. Thecommunication device of claim 1, wherein the control circuit determinesnew reflection coefficient information to initiate a subsequent tuningprocess.
 6. The communication device of claim 5, wherein the impedancetuning process is initially a coarse tuning process, and wherein whenthe new reflection coefficient information has achieved a desired value,the impedance tuning process transitions to a fine tuning process thatadjusts the reactance of the variable reactance element by applyingiterative adjustments to control signals generated by the controlcircuit.
 7. The communication device of claim 6, wherein the fine tuningprocess is configured to compensate for variations in manufacturingtolerances or temperature variations of the matching network.
 8. Thecommunication device of claim 1, wherein control signals generated bythe impedance tuning process are supplied to a plurality of digital toanalog converters that control buffers of the communication device. 9.The communication device of claim 8, wherein the control signalscomprise direct current signals.
 10. The communication device of claim1, wherein the control circuit determines reflection coefficientinformation from the measurements, wherein the reflection coefficientinformation is determined from samples supplied to the control circuitby an analog to digital converter, and wherein control signals aregenerated and directed to a plurality of digital to analog converters.11. The communication device of claim 1, wherein the variable reactanceelement comprises a reactive element controlled by a semiconductordevice.
 12. The communication device of claim 1, wherein the variablereactance element comprises a reactive element controlled by amicro-electro-mechanical systems device.
 13. The communication device ofclaim 1, wherein the variable reactance element comprises a voltagetunable capacitor.
 14. A method, comprising: obtaining, by a controlcircuit of a communication device, measurements via an RF coupler thatsamples forward and reverse waves at a port of a matching network of thecommunication device; and adjusting, by the control circuit, thematching network to perform an impedance tuning process by adjusting areactance of a variable reactance element of the matching networkaccording to the measurements to reduce a return loss.
 15. The method ofclaim 14, further comprising: determining, by the control circuit,reflection coefficient information from the measurements.
 16. The methodof claim 15, wherein the reflection coefficient information isdetermined from samples supplied to the control circuit by an analog todigital converter.
 17. The method of claim 14, wherein a look-up tableis used to perform the impedance tuning process.
 18. The method of claim14, wherein the adjusting of the reactance of the variable reactanceelement of the matching network comprises adjusting a voltage tunablecapacitor.
 19. An apparatus, comprising: a memory that storesinstructions; and a control circuit coupled with the memory, wherein thecontrol circuit, responsive to executing the instructions, performsoperations comprising: obtaining measurements via an RF coupler thatsamples forward and reverse waves at a port of a matching network of acommunication device; and adjusting the matching network to perform animpedance tuning process by adjusting a reactance of a variablereactance element of the matching network according to the measurementsto reduce a return loss.
 20. The apparatus of claim 19, wherein theoperations further comprise determining reflection coefficientinformation from the measurements, wherein the reflection coefficientinformation is determined from samples supplied to the control circuitby an analog to digital converter, and wherein a look-up table is usedto perform the impedance tuning process.